Switching power supply circuit based on ir2153 with voltage regulation. Switching power supply based on IR2153. Electronic Component Ratings

But not one, but four at once. This material will present you with several circuits of switching power supplies made on the popular and reliable IR2153 microcircuit. All these projects were developed by famous user Nem0. Therefore, I will write here on his behalf. All the schematic solutions shown here were personally assembled and tested by the author a couple of years ago.

In general, let's start with the so-called “high-voltage” power supply:

The circuit is traditional, which Nem0 uses in most of its impulse designs. The driver receives power directly from the mains through a resistor. This, in turn, helps to reduce the power dissipated by this resistance, compared to supplying voltage from a 310v circuit. Switching power supply circuit has a smooth voltage switching function, which significantly limits the starting current. The soft start module is powered through capacitor C2, which reduces the mains voltage of 230v.

The power supply provides effective protection to prevent short circuits and peak loads in the secondary power path. The role of the current sensor is performed by a constant resistor R11, and the protection current is adjusted using the trimmer R10. When the current is cut off by the protection, the LED lights up, indicating that the protection has tripped. The output bipolar rectified voltage is +/-70v.

The transformer is made with one primary winding, consisting of fifty turns, and 4 secondary windings, each containing twenty-three turns. The diameter of the copper core and the magnetic circuit of the transformer are calculated depending on the specified power of a specific power supply.

Now consider the following power supply:

This version of the power supply is very similar to the circuit described above, although there are significant differences. The fact is that here the supply voltage to the driver comes from a special winding of the transformer, through a ballast resistor. All other components in the design are almost the same.

The output power of this power supply is determined both by the characteristics of the transformer and the parameters of the IR2153 microcircuit, but also by the life of the diodes in the rectifier. This circuit used KD213A diodes, which have a maximum reverse voltage of 200v and a maximum forward current of 10A. To ensure correct operation of diodes at high currents, they must be installed on a radiator.

The T2 throttle deserves special attention. It is wound on a joint ring magnetic core; if necessary, another core can be used. Winding is done with enamel wire with a cross-section calculated according to the current in the load. Also, the power of the pulse transformer is determined depending on what output power you want to receive. It is very convenient to make calculations of transformers using special computer calculators.

Now the third circuit of a switching power supply based on powerful field-effect transistors IRFP460:

This version of the circuit already has a specific difference compared to previous models. The main differences are that the short circuit and overload protection system is assembled here using a current transformer. And there is one more difference, this is the presence in the circuit of a pair of BD140 pre-output transistors. It is these transistors that make it possible to cut off a large input capacitance of powerful field switches relative to the driver output.

There is also a small difference, this is a voltage suppression resistor related to the soft start module, it is installed in the 230v circuit. In the previous diagram it is located in the +310v power path. In addition, the circuit has an overvoltage limiter that serves to dampen the residual pulse of the transformer. In all other respects, this one no longer has any differences between the above schemes.

DIY PULSE POWER SUPPLY FOR IR2153

Functionally, the IR2153 microcircuits differ only in the Voltage Booster diode installed in the planar housing:

Functional diagram of IR2153

Functional diagram of IR2153D

First, let's look at how the microcircuit itself works, and only then we will decide which power supply to assemble from it. First, let's look at how the generator itself works. The figure below shows a fragment of a resistive divider, three op-amps and an RS trigger:

At the initial moment of time, when the supply voltage has just been applied, capacitor C1 is not charged at all inverting inputs of the op-amp there is a zero, and at the non-inverting inputs there is a positive voltage generated by a resistive divider. As a result, it turns out that the voltage at the inverting inputs is less than at the non-inverting inputs and all three op-amps at their outputs generate a voltage close to the supply voltage, i.e. log unit.
Since the input R (zero setting) on ​​the trigger is inverting, then for it this will be a state in which it does not affect the state of the trigger, but at the input S there will be a log of one, which sets the output of the trigger to also a log of one and a capacitor Ct through resistor R1 will start charging. On the image the voltage across Ct is shown by the blue line,red - voltage at output DA1, green - at output DA2, A pink - at the output of the RS trigger:

As soon as the voltage on Ct exceeds 5 V, a log zero is formed at the DA2 output, and when, continuing to charge Ct, the voltage reaches a value slightly more than 10 volts, a log zero will appear at the DA1 output, which in turn will serve to set the RS trigger to the log zero state. From this moment, Ct will begin to discharge, also through resistor R1, and as soon as the voltage on it becomes slightly less than the set value by dividing the value of 10 V, a log unit will again appear at the output of DA1. When the voltage on the capacitor Ct becomes less than 5 V, a log one will appear at the output of DA2 and turn the RS trigger into the one state and Ct will begin to charge again. Of course, at the inverse output of the RS trigger, the voltage will have opposite logical values.
Thus, at the outputs of the RS trigger, levels of log one and zero are formed that are opposite in phase, but equal in duration:

Since the duration of the control pulses IR2153 depends on the charge-discharge rate of the capacitor Ct, it is necessary to carefully pay attention to flushing the board from flux - there should be no leaks from the terminals of the capacitor or from the printed conductors of the board, since this is fraught with magnetization of the core of the power transformer and failure power transistors.
There are also two more modules in the chip - UV DETECT And LOGIK. The first of them is responsible for starting and stopping the generating process, depending on the supply voltage, and the second generates pulses DEAD TIME, which are necessary to eliminate the through current of the power stage.
Next, the logical levels are separated - one becomes the control upper arm of the half-bridge, and the second the lower one. The difference is that the upper arm is controlled by two field-effect transistors, which, in turn, control the final stage, which is “detached” from the ground and “detached” from the supply voltage. If we consider a simplified circuit diagram for connecting the IR2153, it turns out something like this:

Pins 8, 7 and 6 of the IR2153 microcircuit are outputs VB, HO and VS, respectively, i.e. power supply for the upper side control, the output of the final stage of the upper side control and the negative wire of the upper side control module. Attention should be paid to the fact that at the moment of switching on, the control voltage is present at the Q RS trigger, therefore the low-side power transistor is open. Capacitor C3 is charged through diode VD1, since its lower terminal is connected to the common wire through transistor VT2.
As soon as the RS trigger of the microcircuit changes its state, VT2 closes, and the control voltage at pin 7 of IR2153 opens transistor VT1. At this moment, the voltage at pin 6 of the microcircuit begins to increase, and to keep VT1 open, the voltage at its gate must be greater than at the source. Since the resistance of an open transistor is equal to tenths of an ohm, the voltage at its drain is not much greater than at the source. It turns out that to keep the transistor open, you need a voltage at least 5 volts higher than the supply voltage, and indeed it is - capacitor C3 is charged to 15 volts and it is this that allows you to keep VT1 in the open state, since the energy stored in it during this the moment of time is the supply voltage for the upper arm of the window stage of the microcircuit. Diode VD1 at this point in time does not allow C3 to discharge to the power bus of the microcircuit itself.
As soon as the control pulse at pin 7 ends, transistor VT1 closes and then VT2 opens, which again charges capacitor C3 to a voltage of 15 V.

Quite often, amateurs install an electrolytic capacitor with a capacity of 10 to 100 μF in parallel with capacitor C3, without even delving into the need for this capacitor. The fact is that the microcircuit is capable of operating at frequencies from 10 Hz to 300 kHz and the need for this electrolyte is relevant only up to frequencies of 10 kHz, and then only on condition that the electrolytic capacitor is of the WL or WZ series - technologically they have a small ers and are better known as computer capacitors with inscriptions in gold or silver paint:

For popular conversion frequencies used in the creation of switching power supplies, frequencies are taken above 40 kHz, and sometimes raised to 60-80 kHz, so the relevance of using an electrolyte simply disappears - even a capacitance of 0.22 μF is already enough to open and hold the SPW47N60C3 transistor open , which has a gate capacitance of 6800 pF. To ease the conscience, a 1 µF capacitor is installed, and allowing for the fact that IR2153 cannot switch such powerful transistors directly, the accumulated energy of capacitor C3 is enough to control transistors with a gate capacitance of up to 2000 pF, i.e. all transistors with a maximum current of about 10 A (the list of transistors is below in the table). If you still have doubts, then instead of the recommended 1 µF, use a 4.7 µF ceramic capacitor, but this is pointless:

It would be unfair not to note that the IR2153 microcircuit has analogues, i.e. microcircuits with a similar functional purpose. These are IR2151 and IR2155. For clarity, let’s put the main parameters in a table, and then we’ll figure out which of them is best to prepare:

CHIP	Maximum Driver Voltage	Start supply voltage	Stop supply voltage	Maximum current for charging the gates of power transistors / rise time	Maximum power transistor gate discharge current/fall time	Internal Zener diode voltage
				100 mA / 80...120 nS	210 mA / 40...70 nS
				NOT SPECIFIED / 80...150 nS	NOT SPECIFIED / 45...100 nS
				210 mA / 80...120 nS	420 mA / 40...70 nS

As can be seen from the table, the differences between the microcircuits are not very large - all three have the same shunt zener diode power supply, the start and stop supply voltages are almost the same for all three. The difference lies only in the maximum current of the final stage, which determines which power transistors and at what frequencies the microcircuits can control. Oddly enough, the most hyped IR2153 turned out to be neither fish nor fowl - it does not have a standardized maximum current of the last driver stage, and the rise-fall time is somewhat prolonged. They also differ in cost - IR2153 is the cheapest, but IR2155 is the most expensive.
The generator frequency is the conversion frequency ( no need to divide by 2) for IR2151 and IR2155 is determined by the formulas given below, and the frequency of IR2153 can be determined from the graph:

In order to find out which transistors can be controlled by the IR2151, IR2153 and IR2155 microcircuits, you should know the parameters of these transistors. The greatest interest when connecting a microcircuit and power transistors is the gate energy Qg, since it is this energy that will influence the instantaneous values ​​of the maximum current of the microcircuit drivers, which means a table with transistor parameters will be required. Here SPECIAL Attention should be paid to the manufacturer, since this parameter differs from different manufacturers. This is most clearly seen in the example of the IRFP450 transistor.
I understand perfectly well that for a one-time production of a power supply, ten to twenty transistors are still too much, nevertheless, I posted a link for each type of transistor - I usually buy there. So click, see the prices, compare with retail and the likelihood of buying lefty. Of course, I’m not saying that on Ali there are only honest sellers and all the goods are of the highest quality - there are a lot of scammers everywhere. However, if you order transistors that are produced directly in China, it is much more difficult to run into crap. And it is for this reason that I prefer STP and STW transistors, and I don’t even hesitate to buy them from disassembly, i.e. BOO.

POPULAR TRANSISTORS FOR PULSE POWER SUPPLY
NAME	VOLTAGE			POWER	CAPACITY SHUTTER	Qg (MANUFACTURER)
NETWORK (220 V)
						17...23nC ( ST)
						38...50nC ( ST)
						35...40nC ( ST)
						39...50nC ( ST)
						46nC ( ST)
						50...70nC ( ST)
						75nC ( ST)
						84nC ( ST)
						65nC ( ST)
						46nC ( ST)
						50...70nC ( ST)
						75nC ( ST)
						65nC ( ST)
STP20NM60FP						54nC ( ST)
						150nC(IR) 75nC ( ST)
						150...200nC (IN)
						252...320nC (IN)
						87...117nC ( ST)

I g = Q g / t on = 63 x 10 -9 / 120 x 10 –9 = 0.525 (A) (1)

When the amplitude of the control voltage pulses at the gate is Ug = 15 V, the sum of the output resistance of the driver and the resistance of the limiting resistor should not exceed:

Rmax = U g / I g = 15 / 0.525 = 29 (Ohm) (2)

Let's calculate the output impedance of the driver stage for the IR2155 chip:

R on = U cc / I max = 15V / 210mA = 71.43 ohms
R off = U cc / I max = 15V / 420mA = 33.71 ohms

Taking into account the calculated value according to formula (2) Rmax = 29 Ohm, we come to the conclusion that with the IR2155 driver it is impossible to achieve the specified speed of the IRF840 transistor. If a resistor Rg = 22 Ohm is installed in the gate circuit, the turn-on time of the transistor will be determined as follows:

RE on = R on + R gate, where RE - total resistance, R R gate - resistance installed in the gate circuit of the power transistor = 71.43 + 22 = 93.43 ohms;
I on = U g / RE on, where I on is the opening current, U g - gate control voltage value = 15 / 93.43 = 160mA;
t on = Q g / I on = 63 x 10-9 / 0.16 = 392nS
The shutdown time can be calculated using the same formulas:
RE off = R out + R gate, where RE - total resistance, R out - driver output impedance, R gate - resistance installed in the gate circuit of the power transistor = 36.71 + 22 = 57.71 ohms;
I off = U g / RE off, where I off - opening current, U g - gate control voltage value = 15 / 58 = 259mA;
t off = Q g / I off = 63 x 10-9 / 0.26 = 242nS
To the resulting values ​​it is necessary to add the time of the transistor’s own opening and closing, resulting in the real time t on will be 392 + 40 = 432nS, and t off 242 + 80 = 322nS.
Now all that remains is to make sure that one power transistor has time to close completely before the second one begins to open. To do this, add t on and off getting 432 + 322 = 754 nS, i.e. 0.754 µS. What is it for? The fact is that any of the microcircuits, be it IR2151, or IR2153, or IR2155, has a fixed value DEAD TIME, which is 1.2 µS and does not depend on the frequency of the master oscillator. The datasheet mentions that Deadtime (typ.) 1.2 µs, but it also contains a very confusing drawing from which the conclusion suggests itself that DEAD TIME is 10% of the control pulse duration:

To dispel doubts, the microcircuit was turned on and a two-channel oscilloscope was connected to it:

The power supply was 15 V, and the frequency was 96 kHz. As can be seen from the photograph, with a scan of 1 µS, the duration of the pause is quite a bit more than one division, which exactly corresponds to approximately 1.2 µS. Next we reduce the frequency and see the following:

As can be seen from the photo, at a frequency of 47 kHz, the pause time practically did not change, therefore the sign stating that Deadtime (typ.) 1.2 µs is true.
Since the microcircuits were already working, it was impossible to resist one more experiment - lowering the supply voltage to make sure that the generator frequency would increase. The result is the following picture:

However, expectations were not met - instead of increasing the frequency, it decreased, by less than 2%, which can be generally ignored and noted that the IR2153 microcircuit keeps the frequency quite stable - the supply voltage has changed by more than 30%. It should also be noted that the pause time has increased slightly. This fact is somewhat pleasing - as the control voltage decreases, the opening and closing time of the power transistors increases slightly and increasing the pause in this case will be very useful.
It was also found that UV DETECT copes with its function perfectly - with a further decrease in the supply voltage, the generator stopped, and with an increase, the microcircuit started again.
Now let’s return to our mathematics, based on the results of which we found that with 22 Ohm resistors installed in the gates, the closing and opening time is equal to 0.754 µS for the IRF840 transistor, which is less than the 1.2 µS pause given by the microcircuit itself.
Thus, with the IR2155 microcircuit through 22 Ohm resistors it will be quite normal to control the IRF840, but the IR2151 will most likely have a long life, since to close and open the transistors we needed a current of 259 mA and 160 mA, respectively, and its maximum values ​​are 210 mA and 100 ma. Of course, you can increase the resistance installed in the gates of power transistors, but in this case there is a risk of going beyond the limits DEAD TIME. In order not to engage in fortune telling on coffee grounds, a table was compiled in EXCEL, which you can take. It is assumed that the supply voltage of the microcircuit is 15 V.
To reduce switching noise and slightly reduce the closing time of power transistors in switching power supplies, either the power transistor is shunted with a resistor and capacitor connected in series, or the power transformer itself is shunted with the same chain. This node is called a snubber. The snubber circuit resistor is chosen with a value 5–10 times greater than the drain-source resistance of the field-effect transistor in the open state. The capacitance of the circuit capacitor is determined from the expression:
C = tdt/30 x R
where tdt is the pause time for switching the upper and lower transistors. Based on the fact that the duration of the transient process, equal to 3RC, should be 10 times less than the duration of the dead time value tdt.
Damping delays the opening and closing moments of the field-effect transistor relative to differences in the control voltage across its gate and reduces the rate of change in voltage between the drain and the gate. As a result, the peak values ​​of the flowing current pulses are smaller and their duration is longer. Almost without changing the turn-on time, the damping circuit noticeably reduces the turn-off time of the field-effect transistor and limits the spectrum of generated radio interference.

Now that we've sorted out the theory a little, we can move on to practical schemes.
The simplest switching power supply circuit based on IR2153 is an electronic transformer with a minimum of functions:

The circuit does not have any additional functions, and the secondary bipolar power supply is formed by two rectifiers with a midpoint and a pair of dual Schottky diodes. The capacitance of capacitor C3 is determined at the rate of 1 μF of capacitance per 1 W of load. Capacitors C7 and C8 are of equal capacity and range from 1 µF to 2.2 µF. The power depends on the core used and the maximum current of the power transistors and theoretically can reach 1500 W. However, this is only THEORETICALLY , based on the fact that 155 VAC is applied to the transformer, and the maximum current of the STP10NK60Z reaches 10A. In practice, all datasheets indicate a decrease in the maximum current depending on the temperature of the transistor crystal, and for the STP10NK60Z transistor the maximum current is 10 A at a crystal temperature of 25 degrees Celsius. At a crystal temperature of 100 degrees Celsius, the maximum current is already 5.7 A and we are talking specifically about the temperature of the crystal, and not the heat sink flange, and even more so about the temperature of the radiator.
Therefore, the maximum power should be selected based on the maximum current of the transistor divided by 3 if it is a power supply for a power amplifier and divided by 4 if it is a power supply for a constant load, such as incandescent lamps.
Considering the above, we find that for a power amplifier you can get a switching power supply with a power of 10 / 3 = 3.3A, 3.3A x 155V = 511W. For a constant load we get a power supply 10/4 = 2.5 A, 2.5 A x 155V = 387W. In both cases, 100% efficiency is used, which does not happen in nature. In addition, if we assume that 1 µF of the primary power supply capacity per 1 W of load power, then we will need a capacitor, or capacitors with a capacity of 1500 µF, and such a capacitance must be charged through soft start systems.
A switching power supply with overload protection and soft start via secondary power is presented in the following diagram:

First of all, this power supply has overload protection made on the current transformer. You can read details about calculating a current transformer. However, in the vast majority of cases, a ferrite ring with a diameter of 12...16 mm, on which about 60...80 turns are wound in two wires, is quite sufficient. Diameter 0.1...0.15 mm. Then the beginning of one winding is connected to the ends of the second. This is the secondary winding. The primary winding contains one or two, sometimes one and a half turns are more convenient.
Also in the circuit, the values ​​of resistor R4 and R6 are reduced in order to expand the range of the primary supply voltage (180...240V). In order not to overload the zener diode installed in the microcircuit, the circuit has a separate zener diode with a power of 1.3 W at 15 V.
In addition, a soft start for secondary power was introduced into the power supply, which made it possible to increase the capacitance of the secondary power filters to 1000 µF at an output voltage of ±80 V. Without this system, the power supply entered protection at the moment of switching on. The principle of operation of the protection is based on the operation of IR2153 at an increased frequency at the moment of switching on. This causes losses in the transformer and it is not able to deliver maximum power to the load. As soon as generation begins through the divider R8-R9, the voltage supplied to the transformer reaches the detector VD5 and VD7 and charging of the capacitor C7 begins. As soon as the voltage becomes sufficient to open VT1, C3 is connected to the frequency-setting chain of the microcircuit and the microcircuit reaches the operating frequency.
Additional inductances for the primary and secondary voltages have also been introduced. Inductance on the primary power supply reduces interference created by the power supply and going into the 220V network, and on the secondary power supply it reduces RF ripple on the load.
In this version there are two additional secondary supplies. The first is intended to power a twelve-volt computer cooler, and the second is to power the preliminary stages of a power amplifier.
Another sub-option of the circuit is a switching power supply with a unipolar output voltage:

Of course, the secondary winding is designed for the voltage that is required. The power supply can be soldered on the same board without mounting elements that are not on the diagram.

The next version of the switching power supply is capable of delivering about 1500 W to the load and contains soft start systems for both primary and secondary power, has overload protection and voltage for the forced cooling cooler. The problem of controlling powerful power transistors is solved by using emitter followers on transistors VT1 and VT2, which discharge the gate capacitance of powerful transistors through themselves:

Such forcing of the closing of power transistors allows the use of quite powerful specimens, such as IRFPS37N50A, SPW35N60C3, not to mention IRFP360 and IRFP460.
At the moment of switching on, the voltage is supplied to the primary power diode bridge through resistor R1, since the contacts of relay K1 are open. Next, the voltage is supplied through R5 to the microcircuit and through R11 and R12 to the output of the relay winding. However, the voltage increases gradually - C10 has a fairly large capacity. From the second winding of the relay, voltage is supplied to the zener diode and thyristor VS2. As soon as the voltage reaches 13 V, it will be enough to pass through the 12-volt zener diode to open VS2. Here it should be recalled that IR2155 starts with a supply voltage of approximately 9 V, therefore, at the time of opening, VS2 will already generate control pulses through IR2155, only they will enter the primary winding through resistor R17 and capacitor C14, since the second group of contacts of relay K1 is also open . This will significantly limit the charging current of the secondary power filter capacitors. As soon as the thyristor VS2 opens, voltage will be applied to the relay winding and both contact groups will close. The first will bypass the current-limiting resistor R1, and the second - R17 and C14.
The power transformer has a service winding and a rectifier on diodes VD10 and VD11 from which the relay will be powered, as well as additional power supply to the microcircuit. R14 serves to limit the forced cooling fan current.
Thyristors used VS1 and VS2 - MCR100-8 or similar in TO-92 housing
Well, at the end of this page, another circuit is still on the same IR2155, but this time it will act as a voltage stabilizer:

As in the previous version, the power transistors are closed by bipolars VT4 and VT5. The circuit is equipped with a soft start of the secondary voltage on VT1. The start is made from the vehicle’s on-board power supply and then the power is supplied by a stabilized voltage of 15 V, vortexed by diodes VD8, VD9, resistor R10 and zener diode VD6.
There is another rather interesting element in this circuit - tC. This is heatsink overheat protection that can be used with almost any converter. It was not possible to find an unambiguous name; in common parlance it is a self-restoring thermal fuse; in price lists it is usually designated KSD301. It is used in many household electrical appliances as a protective or temperature-regulating element, since they are produced with different response temperatures. This fuse looks like this:

As soon as the radiator temperature reaches the fuse cut-off limit, the control voltage from the REM point will be removed and the converter will turn off. After the temperature drops by 5-10 degrees, the fuse will be restored and supply control voltage and the converter will start again. The same thermal fuse, or thermal relay, can also be used in network power supplies by monitoring the temperature of the radiator and turning off the power, preferably low-voltage, going to the microcircuit - the thermal relay will work longer this way. You can buy KSD301.
VD4, VD5 - fast diodes from the SF16, HER106, etc. series.
Overload protection can be introduced into the circuit, but during its development the main emphasis was on miniaturization - even the soft start unit was a big question.
The manufacture of winding parts and printed circuit boards are described on the following pages of the article.

Well, at the end of the day there are several circuits of switching power supplies found on the Internet.
Scheme No. 6 taken from the SOLDERING IRON website:

In the next power supply on the self-clocked driver IR2153, the capacitance of the boost capacitor is reduced to a minimum of 0.22 μF (C10). The microcircuit is powered from an artificial midpoint of the power transformer, which is not important. There is no overload protection; the shape of the voltage supplied to the power transformer is slightly corrected by the inductance L1:

While selecting diagrams for this article, I came across this one. The idea is to use two IR2153 in a bridge converter. The author's idea is quite clear - the output of the RS trigger is fed to the input Ct and, according to the logic, control pulses of opposite phase should be generated at the outputs of the slave microcircuit.
The idea intrigued me and an investigative experiment was carried out on the topic of testing its functionality. It was not possible to obtain stable control pulses at the outputs of IC2 - either the upper driver or the lower one was working. In addition, the pause phase changed DEAD TIME, on one microcircuit relative to another, which will significantly reduce the efficiency and the idea was forced to be abandoned.

The distinctive feature of the next power supply on the IR2153 is that if it works, then this work is akin to a powder keg. First of all, the additional winding on the power transformer to power the IR2153 itself caught my eye. However, there is no current-limiting resistor after diodes D3 and D6, which means that the fifteen-volt zener diode located inside the microcircuit will be VERY heavily loaded. One can only guess what will happen if it overheats and undergoes thermal breakdown.
The overload protection on VT3 bypasses the time setting capacitor C13, which is quite acceptable.

The last acceptable version of the power source circuit on the IR2153 does not represent anything unique. True, for some reason the author too reduced the resistance of the resistors in the gates of the power transistors and installed zener diodes D2 and D3, the purpose of which is not very clear. In addition, the capacitance C11 is too small, although it is possible that we are talking about a resonant converter.

There is another option for a switching power supply using IR2155 and specifically for controlling a bridge converter. But there the microcircuit controls power transistors through an additional driver and matching transformer, and we are talking about induction melting of metals, so this option deserves a separate page, and everyone who understood at least half of what they read should go to the page with PRINTED BOARDS.

VIDEO INSTRUCTIONS FOR SELF ASSEMBLY
SWITCH POWER SUPPLY BASED ON IR2153 OR IR2155

A few words about the manufacture of pulse transformers:

How to determine the number of turns without knowing the grade of ferrite:

Good day everyone! I’m looking at diagrams on the Internet of switching power supplies and... And I don’t understand! Perhaps the authors don’t read the “Datasheet” for components, or are they specifically discouraged from assembling a UPS??? . Let's look at the description of IR2153: "an improved version of IR2153 -2155, the list of improvements comes down to protection from interference... We read: the recommended load capacitance is 1000 pF, power 0.650 W (short-term)! So this is the data on IR2151!!! And so we have: IR2153 can control keys with a capacitive load of 1n=1000 pf! Look at the "datasheet" of the keys. IR740 - 1450 pf. One and a half times higher than the recommended one. Now the voltage. The recommended maximum voltage of the keys is 600 v (v)! And the keys have 400 v. Well, yes, this more than 310 V! However, everyone who has come across industrial UPS circuits is well aware that switches are placed at a voltage of at least 600 V. Only in Chinese circuits sometimes burnt-out ones at 500 V appear. I hope I explained it clearly?! As for the switch current and resistance key in the open state. This has little effect on the power of the UPS. Let me explain. For a switching power supply, the current is limited by passing through the load and, as a rule, in a pulse does not exceed 2-3 A. In a pulse! We look at the “datasheet” of the keys and see: at a crystal temperature of 100 gr. current with a large margin for the IR740. However, in this case this is a minus for the key! The higher the switch current, the longer the switching time (see the graph there) and, of course, the lower the pulse slope, which means the efficiency is less than the maximum (75%). Accordingly, this key will work, but poorly!!! As a result of the above: this combination leads to burnout of both the keys and the driver! Anyone who wants to repeat this scheme is doomed to a handful of burnt parts! I am wrong? Read the comments on similar diagrams. The question follows: you are so smart, so what do you recommend? I advise everyone who wants to have a simple UPS assembly to take the diagram from the description and recommendation of the IR Company - IR2153 driver with switches for a current of 4-5 A and max. voltage 600-900 V with a control electrode capacitance of no more than 1000 pF. Example STP5NK600C and similar MOSFET triodes. Now about the resistance in the open state for the key: indeed, the greater it is, the stronger the heating of the key. Some will say less efficiency. In this case, the efficiency is not 100% and the effect of resistance is very small. So what affects efficiency? The efficiency is affected by the UPS circuit itself; for an efficiency of up to 94%, we assemble a resonant UPS. Efficiency up to 75% - with the right keys on the IR2153!. Is this efficiency not enough for you? Hm. What about a pulse transformer? How will it limit efficiency? Has anyone already counted? Losses at frequencies above 50 kHz increase significantly, although losses up to 50 kHz are not zero. Let's look at industrial circuits: winding pulse transformers is a very capricious task; two equally wound transformers have different inductances! What is this? And this is what it is! Each IT has its own optimal operating frequency. How do you like this? That's it - read on and look at the UPS diagrams for TVs, powerful amplifiers, and other factory electrical appliances. Good luck to you!

Power supply

Switching amplifier power supply for IR2151, IR2153

Switching power supplies are the most efficient class of secondary power supplies. They are characterized by compact size, high reliability and efficiency. The only disadvantages include the creation of high-frequency interference and the complexity of design/implementation.

All pulse power banks are a kind of inverters (systems that generate alternating voltage at the high-frequency output from the rectified voltage at the input).
The complexity of such systems is not even in first rectifying the input mains voltage, or subsequently converting the output high-frequency signal into a constant one, but in the feedback, which allows you to effectively stabilize the output voltage.

Particularly complex here is the process of controlling high-level output voltages. Very often the control unit is powered from low voltage, which creates the need to coordinate levels.

Drivers IR2151, IR2153

In order to control independently (or dependently, but with a special pause that prevents simultaneous opening of the keys) the channels of the upper and lower keys, self-clocked half-bridge drivers are used, such as IR2151 or IR2153 (the latter chip is an improved version of the original IR2151, both are interchangeable).

There are numerous modifications of these circuits and analogues from other manufacturers.

A typical driver circuit with transistors looks like this.

Rice. 1. Driver connection circuit with transistors

The package type can be PDIP or SOIC (the difference is in the picture below).

Rice. 2. Package type PDIP and SOIC

The modification with the letter D at the end assumes the presence of an additional boost diode.

The differences between the IR2151 / 2153 / 2155 microcircuits in parameters can be seen in the table below.

Table

UPS on IR2153 - the simplest option

The schematic diagram itself looks like this.

Rice. 3. Schematic diagram of the UPS

At the output, you can get bipolar power (implemented by rectifiers with a midpoint).

The power of the power supply can be increased by changing the capacitance parameters of capacitor C3 (calculated as 1:1 - 1 µF is required for 1 W of load).

In theory, the output power can be increased to 1.5 kW (although capacitors of such a capacity will require a soft start system).

With the configuration indicated in the circuit diagram, an output current of 3.3A (up to 511 V) is achieved when used in power amplifiers, or 2.5A (387 V) when connecting a constant load.

UPS with overload protection

The scheme itself.

Rice. 4. UPS circuit with overload protection

This power supply provides a system for switching to the operating frequency, eliminating inrush current surges (soft start), as well as simple protection against RF interference (at the input and output of the inductor).

UPS up to 1.5 kW

The circuit below can handle high-power power transistors such as SPW35N60C3, IRFP460, etc.

Rice. 5. UPS diagram with power up to 1.5 kW

Powerful VT4 and VT5 are controlled through emitter followers on VT2 and VT1.

Amplifier power supply on a transformer from a computer power supply

It often happens that there is practically no need to buy components; they can sit and gather dust as part of equipment that has not been used for a long time, for example, in a PC system unit somewhere in the basement or on the balcony.

Below is one of the fairly simple, but no less efficient UPS circuits for an amplifier.

Alexander / 04/24/2019 - 08:24
in Fig. 6 there is an error: there is no capacitor in the output transformer circuit